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| HV9912 Preliminary Switch-mode LED Driver IC With High Current Accuracy and Hiccup Mode Protection Features Switch mode controller for single switch drivers Buck Boost Buck-boost and SEPIC Works with high side current sensors Closed loop control of output current High PWM dimming ratio Internal 90V linear regulator (can be extended using external zener diodes) Internal 2% voltage reference (0C < TA < 85C) Constant frequency operation Programmable slope compensation Linear & PWM dimming +0.2A/-0.4A gate drive Hiccup Mode Protection for both short circuit and open circuit conditions Synchronization capability Pin compatible with HV9911 General Description The HV9912 is a current mode control LED driver IC designed to control single switch PWM converters (buck, boost, buck-boost or SEPIC) in a constant frequency mode. The controller uses a peak current-mode control scheme with programmable slope compensation and includes an internal transconductance amplifier to control the output current in closed loop enabling high output current accuracy (in the case of buck and buck-boost converters, the output current can be sensed using a high side current sensor like the HV7800). In the constant frequency mode, multiple HV9912 ICs can by synchronized to each other or to an external clock using the SYNC pin. Programmable MOSFET current limit enables current limiting during input under voltage and output overload conditions. The IC also includes a 0.2A source and 0.4A sink gate driver that makes the HV9912 suitable for high power applications. An internal 90V linear regulator powers the IC eliminating the need for a separate power supply for the IC. The IC also provides a FAULT output, which can be used to disconnect the LEDs in case of a fault condition using an external disconnect FET. HV9912 also provides a TTL compatible, low-frequency PWM dimming input that can accept an external control signal with a duty ratio of 0-100% and a frequency of up to a few kilohertz. The HV9912 includes hiccup protection from both short and open circuits, with automatic recovery after the fault condition is cleared. The HV9912 is a pin compatible replacement to Supertex's HV9911. It is compatible with existing HV9911 designs which have an input voltage of less than 90V by changing ROVP1, ROVP, and RT. Applications LED backlight applications General LED lighting applications Battery powered LED lamps Typical Application Circuit - Boost D2 VIN CIN 1 VIN GATE 3 L1 Q1 D1 ROVP1 CO CDD 2 VDD CS 5 RSC RCS ROVP2 4 GND OVP 12 RSLOPE 6 SC FAULT 11 Q2 RT 7 HV9912 RT FDBK 16 CREF Cc 10 REF COMP 14 Rs 9 CLIM PWMD 13 RL1 RL2 15 IREF SYNC 8 RR1 RR2 HV9912 Preliminary Ordering Information DEVICE HV9912 -G indicates package is RoHS compliant (`Green') Pin Configuration Package Options 16-Lead SOIC HV9912NG-G 3 4 5 6 7 8 GATE COMP 1 2 VIN FDBK 16 15 14 13 12 11 10 9 VDD IREF GND PWMD HV9912 CS OVP FAULT SC RT REF CLIM Absolute Maximum Ratings Parameter VIN to GND VDD to GND CS to GND PWMD to GND GATE to GND All other pins to GND Value -0.5V to +100V -0.3V to +13.5V -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V) 1200mW 82OC/W +125C -65C to +150C SYNC 16-Lead SOIC (NG) (top view) Product Marking Top Marking HV9912NG YWW LLLLLLLL Continuous Power Dissipation (TA = +25C) Thermal impedance (ja) Junction temperature Storage temperature range Bottom Marking CCCCCCCCC AAA Y = Last Digit of Year Sealed WW = Week Sealed L = Lot Number C = Country of Origin* A = Assembler ID* = "Green" Packaging *May be part of top marking 16-Lead SOIC (NG) Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Electrical Characteristics (The * denotes the specifications which apply over the full operating ambient temperature range of 0OC < TA < +85OC, otherwise the specifications are at TA = 25OC. VIN = 12V, unless otherwise noted.) Sym Input VINDC IINSD Parameter Min Typ Max Units Conditions Input DC supply voltage range Shut-down mode supply current * * (1) - 90 1.5 V mA DC input voltage PWMD connected to GND - Internal Regulator VDD UVLORISE UVLOHYST Internally regulated voltage VDD undervoltage lockout threshold VDD undervoltage lockout hysteresis * 7.25 6.5 7.75 500 8.25 7.0 V V mV VIN = 9.0 - 90V, PWMD connected to GND VDD rising VDD falling 2 HV9912 Preliminary Electrical Characteristics (cont.) (The * denotes the specifications which apply over the full operating ambient temperature range of 0OC < TA < +85OC, otherwise the specifications are at TA = 25OC. VIN = 12V, unless otherwise noted.) Sym Reference VREF Parameter Min Typ Max Units Conditions REF pin voltage * 1.225 1.25 1.275 V REF bypassed with a 0.1F capacitor to GND; IREF = 0; PWMD = GND REF bypassed with a 0.1F capacitor to GND; IREF = 0; VDD = 7.25 - 12V; PWMD = GND REF bypassed with a 0.1F capacitor to GND; IREF = 0-500A; PWMD = GND ----VPWMD = 5.0V VGATE = 0V VGATE = VDD CGATE = 1.0nF CGATE = 1.0nF VREFLINE Line regulation of reference voltage Load regulation of reference voltage 0 - 20 mV VREFLOAD 0 - 10 mV PWM Dimming VPWMD(lo) VPWMD(hi) RPWMD GATE ISOURCE ISINK TRISE TFALL GATE short circuit current GATE sinking current GATE output rise time GATE output fall time 0.2 0.4 50 25 - PWMD input low voltage PWMD input high voltage PWMD pull-down resistance * * 2.0 50 100 0.8 150 V V k A A ns ns 85 45 Over Voltage Protection VOVP, RISING VOVP, HYST TBLANK TDELAY1 TDELAY2 VOFFSET GB AV VCM VO Gm VOFFSET IBIAS Over voltage rising trip point Over voltage Hysteresis 4.75 5.00 0.50 5.25 V V OVP rising OVP falling Current Sense Leading edge blanking Delay to output of COMP comparator Delay to output of CLIMIT comparator Comparator offset voltage * 100 -10 250 200 200 10 ns ns ns mV --COMP = VDD ; CLIM = REF; CSENSE = 0 to 600mV step COMP = VDD ; CLIM = 300mV; CSENSE = 0 to 400mV step --- Internal Transconductance Opamp Gain bandwidth product Open loop DC gain Input common-mode range Output voltage range Transconductance Input offset voltage Input bias current # # # # 60 -0.3 0.7 450 -5.0 1.0 550 0.5 3.0 6.75 650 5.0 1.0 MHz dB V V A/V mV nA 75pF capacitance at OP pin Output open ----------- 3 HV9912 Preliminary Electrical Characteristics (cont.) (Over recommended operating conditions. VIN = 24V, TA = 25C, unless otherwise specified) Sym Oscillator fOSC1 fOSC2 DMAX VSYNCH VSYNCL IOUTSYNC GSC VOMIN Parameter Min Typ Max Units Conditions Oscillator frequency Oscillator frequency Maximum duty cycle SYNC input high SYNC input low SYNC output current * * 99 510 87 2.0 - 106 580 18 118 650 93 0.8 - kHz kHz % V V A RT = 500k RT = 96k --------- Output Short Circuit Gain for short circuit comparator Minimum output voltage of the gain stage Propagation time for short circuit detection Fault output rise time Fault output fall time Blanking time Current source at COMP pin used for hiccup mode protection Current sourced out of SC pin Internal current mirror ratio * 0 1.8 * 500 * 1.9 0.125 2.0 2.1 0.25 V V --IREF = GND PWMD = VDD, IREF = 400mA; FDBK step from 0 to 900mV; FAULT goes from high to low 330pF capacitor at FAULT pin 330pF capacitor at FAULT pin ----- TOFF TRISE,FAULT TFALL,FAULT TBLANK,SC IHICCUP - 5.0 250 300 300 700 - ns ns ns ns A Slope Compensation ISLOPE GSLOPE 2.0 100 2.2 A --ISLOPE = 50A ; RSC = 1.0k Notes: (1) See Application Information for Minimum Input Voltage. * The specifications which apply over the full operating temperature range at 0OC < TA < +85OC are guaranteed by design and characterization. # Denotes specifications guaranteed by design. 4 HV9912 Preliminary Functional Block Diagram VIN Linear Regulator Vbg REF VDD 5.60/6.10V + _ SS CLIM Blanking CS 1:2 TBLANK _ + POR GATE ramp R Q SS FAULT SC POR COMP Hiccup/Dimming Block PWMD FDBK IREF _ + GM One Shot RT Functional Description Power Topology The HV9912 is a switch-mode converter LED driver designed to control a continuous conduction mode buck or boost in a constant frequency (or constant off-time) mode. The IC includes an internal linear regulator, which operates from input voltages up to 90V eliminating the need for an external power supply for the IC. The IC includes features typically required in LED drivers like open LED protection, output short circuit protection, linear and PWM dimming, programmable input current limiting and accurate control of the LED current. A high current gate drive output enables the controller to be used in high power converters. The HV9912 is an enhanced version of the HV9911 with hysteretic over-voltage protection and hiccup mode short circuit protection. The IC includes a blanking network controlled by the PWMD input to prevent the short circuit protection from triggering prematurely during PWM dimming due to the parasitic capacitance of the LED string. It also allows the IREF pin to be pulled all the way down to GND without triggering the short circuit protection. It is a pin compatible replacement to the HV9911. Linear Regulator The HV9912 can be powered directly from its VIN pin that withstands a voltage up to 90V. When a voltage is applied at the VIN pin, the HV9912 tries to maintain a constant 7.75V (typ) at the VDD pin. The regulator also has a built in under- + _ S OVD SCD 5V rising 4.5V falling OVPD OVP 13R R TBLANK,SC SS SCD SYNC 2 PWMD PWMD GND 5 HV9912 Preliminary voltage lockout which shuts off the IC if the voltage at the VDD pin falls below the UVLO threshold. The VDD pin must be bypassed by a low ESR capacitor (0.1F) to provide a low impedance path for the high frequency current of the output gate driver. The input current drawn from the VIN pin is a sum of the 1.5mA current drawn by the internal circuit and the current drawn by the gate driver (which in turn depends on the switching frequency and the gate charge of the external FET). IIN = 1.5mA + (QG x fS) (Eqn. 1) In this case, the gate drive draws too much current and VINSTART is less than VINSTOP. In such cases, the IC will oscillate between ON and OFF if the input voltage is between the start and stop voltages. In these circumstances, it is recommended that the input voltage be kept higher than VINSTOP. Reference HV9912 includes a 2% accurate, 1.25V reference, which can be used as the reference for the output current as well as to set the switch current limit. The reference is buffered so that it can deliver a maximum of 500A external current to drive the external circuitry. The reference should be bypassed with at least a 10nF low ESR capacitor. Note: In order to avoid abnormal start-up conditions, the bypass capacitor at the REF pin should not exceed 0.1F. Oscillator Connecting a resistor between RT and GND will program the time period. In both cases, resistor RT sets the current which charges an internal oscillator capacitor. The capacitor voltage ramps up linearly and when the voltage increases beyond the internal set voltage, a comparator triggers the SET input of the internal SR flip-flop. This starts the next switching cycle. The time period of the oscillator can be computed as: TS RT x 18pF (Eqn. 3) In the above equation, fs is the switching frequency and QG is the gate charge of the external FET (which can be obtained from the datasheet of the FET). Minimum Input Voltage at VIN pin The minimum input voltage at which the converter will start and stop depends on the minimum voltage drop required for the linear regulator. The internal linear regulator will regulate the voltage at the VDD pin when VIN is between 9V and 90V. However, when VIN is less than 9V, the converter will still function as long as VDD is greater than the under voltage lockout. Thus, the converter might be able to start at lower than 9V. The start/stop voltages at the VIN pin can be determined using the minimum voltage drop across the linear regulator as a function of the current drawn. This data is shown in Fig. 1 for ambient temperatures of 25C and 85C. Assume an ambient temperature of 85OC. Assuming the IC is driving a 15nC gate charge FET at 200kHz, the total input current is estimated to be 4.5mA (using Eqn. 1). At this input current, the minimum voltage drop from Fig. 1 can be approximately estimated to be VDROP = 1.25V. However, before the IC starts switching the current drawn will be 1.5mA. At this current level, the voltage drop is approximately VDROP1 = 0.3V. Thus, the start/stop VIN voltages can be computed to be: VINSTART = UVLOMAX + VDROP1 (Eqn. 2) = 7.0V + 0.3V = 7.3V VINSTOP = UVLOMAX - UVLO + VDROP = 7.0V - 0.5V + 1.25V = 7.75V Fig. 1 Headroom vs Input Current Minimum Voltage Drop vs. IIN 3 Synchronization The SYNC pin is an input/output (I/O) port to a fault tolerant peer-to-peer and/or master clock synchronization circuit. For synchronization, the SYNC pins of multiple HV9912 based converters can be connected together and may also be connected to the open drain output of a master clock. When connected in this manner, the oscillators will lock to the device with the highest operating frequency. When synchronizing multiple ICs, it is recommended that the same timing resistor be (corresponding to the switching frequency) be used in all the HV9912 circuits. In rare occasions, given the length of the connecting lines for the SYNC pins, a resistor between SYNC and GND may be required to damp any ringing due to parasitic capacitances. It is recommended that the resistor chosen be greater than 300k. When synchronized in this manner, a permanent HIGH or LOW condition on the SYNC pin will result in a loss of synchronization, but the HV9912 based converters will continue to operate at their individually set operating frequency. Since loss of synchronization will not result in total system failure, the SYNC pin is considered fault tolerant. Minimum Voltage Drop (V) 2.5 2 1.5 TA = 85OC TA = 25OC 1 0.5 0 0 2 4 IIN (mA) 6 8 10 6 HV9912 Preliminary Slope Compensation For continuous conduction mode converters operating in the constant frequency mode, slope compensation becomes necessary to ensure stability of the peak current mode controller, if the operating duty cycle is greater than 0.5. Choosing a slope compensation which is one half of the down slope of the inductor current ensures that the converter will be stable for all duty cycles. Slope compensation can be programmed by two resistors RSLOPE and RSC. Assuming a down slope of DS (A/s) for the inductor current, the slope compensation resistors can be computed as: RSC = RSLOPE x DS x 106 x TS x RCS 10 where RCS is the current sense resistor which senses the switching FET current. Note: The maximum current that can be sourced out of the SC pin is limited to 100A. This limits the minimum value of the RSLOPE resistor to 25k. If the equation for slope compensation produces a value of RSLOPE less than this value, then RSC would have to be reduced accordingly. It is recommended that RSLOPE be chosen in the range of 25k - 50k. Current Sense The current sense input of the HV9912 includes a built in 100ns (minimum) blanking time to prevent spurious turn off due to the initial current spike when the FET turns on. The HV9912 includes two high-speed comparators - one is used during normal operation and the other is used to limit the maximum input current during input under voltage or overload conditions. The IC includes an internal resistor divider network, which steps down the voltage at the COMP pin by a factor of 15. This stepped-down voltage is given to one of the comparators as the current reference. The reference to the other comparator, which acts to limit the maximum inductor current, is given externally. It is recommended that the sense resistor RCS be chosen so as to provide about 250mV current sense signal. Current Limit Current limit has to be set by a resistor divider from the 1.25V reference available on the IC. Assuming a maximum operating inductor current IPK (including the ripple current), the maximum voltage at the CLIM pin can be set as: VCLIM 1.2 x IPK x RCS + 5 x RCS x 0.9 RSLOPE (Eqn. 5) (Eqn. 4) Note that this equation assumes a current limit at 120% of the maximum input current. Also, if VCLIM is greater than 450mV, the saturation of the internal opamp will determine the limit on the input current rather than the CLIM pin. In such a case, the sense resistor RCS should be reduced till VCLIM reduces below 550mV. It is recommended that no capacitor be connected between CLIM and GND. Internal 1MHz Transconductance Amplifier HV9912 includes a built in 1MHz transconductance amplifier, with tri-state output, which can be used to close the feedback loop. The output current sense signal is connected to the FDBK pin and the current reference is connected to the IREF pin. The output of the opamp is controlled by the signal applied to the PWMD pin. When PWMD is high, the output of the opamp is connected to the COMP pin. When PWMD is low, the output is left open. This enables the integrating capacitor to hold the charge when the PWMD signal has turned off the gate drive. When the IC is enabled, the voltage on the integrating capacitor will force the converter into steady state almost instantaneously. The output of the opamp is buffered and connected to the current sense comparator using a 15:1 divider. The buffer helps to prevent the integrator capacitor from discharging during the PWM dimming state. PWM Dimming PWM dimming can be achieved by driving the PWMD pin with a TTL compatible square wave source. The PWM signal is connected internally to the three different nodes - the transconductance amplifier, the FLT output and the GATE output. When the PWMD signal is high, the GATE and FLT pins are enabled and the output of the transconductance opamp is connected to the external compensation network. Thus, the internal amplifier controls the output current. When the PWMD signal goes low, the output of the transconductance amplifier is disconnected from the compensation network. Thus, the integrating capacitor maintains the voltage across it. The GATE is disabled, so the converter stops switching and the FLT pin goes low, turning off the disconnect switch. The output capacitor of the converter determines the PWM dimming response of the converter, since it has to get charged and discharged whenever the PWMD signal goes high or low. In the case of a buck converter, since the inductor current is continuous, a very small capacitor is used across the LEDs. This minimizes the effect of the capacitor on the PWM dimming response of the converter. However, in the case of a boost converter, the output current is dis- 7 HV9912 Preliminary continuous and a very large output capacitor is required to reduce the ripple in the LED current. Thus, this capacitor will have a significant impact on the PWM dimming response. By turning off the disconnect switch when PWMD goes low, the output capacitor is prevented from being discharged and thus the PWM dimming response of the boost converter is greatly improved. Note that in case of continuous conduction mode boost converters, disconnecting the capacitor might cause a sudden spike in the capacitor voltage as the energy in the inductor is dumped into the capacitor. This increase in the capacitor voltage might cause the OVP comparator to trip if the OVP point is set too close to the maximum operating voltage. Thus, either the capacitor has to larger to absorb this energy without increasing the capacitor voltage significantly or the OVP set point has to be increased. False Triggering of the Short Circuit Comparator During PWM Dimming During PWM dimming, the parasitic capacitance of the LED string causes a spike in the output current when the disconnect FET is turned on. With the HV9911, this parasitic spike in the output current caused the IC to falsely detect an over current condition and shut down. To prevent this false shutdown, an R-C filter was used at the FDBK pin to filter this spike. To prevent these false triggering in the HV9912, there is a built-in 500ns blanking network for the short circuit comparator, which eliminates the need for the external R-C low pass filter. This blanking network is activated when the PWMD input goes high. Thus, the short circuit comparator will not see the spike in the LED current during the PWM Dimming turn-on transition. Once the blanking timer is completed, the short circuit comparator will start monitoring the output current. Thus, the total delay time for detecting a short circuit will depend on the condition of the PWMD input. If the output short circuit exists before the PWMD signal goes high, the total detection time will be: COMP the COMP pin is disconnected from the GM amplifier and the GATE and FLT pins are pulled low disabling the LED driver. When the fault has cleared, a 5A current source is activated which pulls the COMP network up to 5V. Once the voltage at the COMP network reaches 5V, the 5A sourcing current is disconnected and a 5A sinking current is activated which pulls the COMP pin low. When the voltage at the COMP pin reaches 1V, the sinking current is disconnected and the Gm amplifier is reconnected to the COMP pin. The FLT pin goes high and the GATE pin is now allowed to switch. The closed loop control then takes over the control of the LED current. Startup Condition The startup waveforms are shown in Fig. 2. Assuming a pole-zero R-C network at the COMP pin (series combination of RZ and CZ in parallel with CC), the start-up delay time can be approximately computed as tdelay tPOR + ( CC + CZ) x 9V 5A (Eqn. 8) This equation assumes that the voltage drop across RZ can be neglected compared to the voltage swing at the COMP pin, which is true in most of the cases (RZ < 100k). The POR time (tPOR) for the HV9912 is 10s. VIN POR Pull-up with 5 Pull-down with 5 Gm c ontrol tdetect1 = tblank,SC(max)+ tdelay(max) 700 + 250 950ns(max) (Eqn. 6) 5V If the short circuit occurs when the PWMD signal is already high, the time to detect will be: tdetect1 = tdelay(max) 250ns(max) (Eqn. 7) 1V tPOR FLT tdelay Hiccup Timer The HV9912 reuses the compensation network on the COMP pin to create a timer which is activated upon startup or when a detected fault has been cleared. When a fault is detected (either open circuit or short circuit) or upon startup, Fig. 2 Waveforms During Startup 8 HV9912 Preliminary FAULT Condition In the case of a fault condition (either open circuit or short circuit), the same sequence is repeated with the only difference being that the COMP pin voltage does not start from zero, but rather from its steady-state condition. Short Circuit Protection When a short circuit condition is detected (output current becomes higher than twice the steady state current), the GATE and FLT outputs are pulled low. As soon as the disconnect FET is turned off, the output current goes to zero and the short circuit condition disappears. At this time, the hiccup timer is started (Fig. 3). Once the timing is complete, the converter attempts to restart. If the fault condition still persists, the converter shuts down and goes through the cycle again. If the fault condition is cleared (due to a momentary output short) the converter will start regulating the output current normally. This allows the LED driver to recover from accidental shorts without having to reset the IC. The hiccup time will depend on the steady state voltage of the COMP pin (VCOMP). This is typically in the range of 3V 4V. The hiccup time can be approximately computed as: tHICCUP (CC + CZ) x 9V - VCOMP 5A Output Current Short Circuit occurs When the load is disconnected in a boost converter, the output voltage rises as the output capacitor starts charging. When the output voltage reaches the OVP rising threshold, the HV9912 detects an over voltage condition and turns off the converter. The converter is turned back on only when the output voltage falls below the falling OVP threshold (which is 10% lower than the rising threshold). This time is mostly dictated by the R-C time constant of the output capacitor Co and the resistor network used to sense over voltage (ROVP1 + ROVP2). In case of a persistent open circuit condition, this cycle keeps repeating maintaining the output voltage within a 10% band. In most designs, the lower threshold voltage of the over voltage protection when the converter will be turned on will be more than the LED string voltage. Thus, when the LED load is reconnected to the output of the converter, the voltage differential between the actual output voltage and the LED string voltage will cause a spike in the output current when the FLT signal goes high. This causes a short circuit to be detected and the HV9912 will go into short circuit protection. This behavior continues till the output voltage becomes lower than the LED string voltage, at which point no fault will be detected and normal operation of the circuit will commence (Fig. 4). The various delay times can be computed as follows: tRC 0.1 x (ROVP1 + ROVP2) x CO (Eqn. 10) (Eqn. 9) Normal Operation Res umes tHICCUP1 (CC + CZ) x 9V - VCOMP 5A tHICCUP2-n (CC + CZ) x 9V 5A (Eqn. 11) FLT Hiccup Time (Eqn. 12) Note that the number of hiccup cycles might be more than two. COMP 5V 1V thiccup Fig. 3 Short Circuit Protection Over Voltage Protection The HV9912 provides hysteretic over voltage protection allowing the IC to recover in case the LED load is disconnected momentarily. 9 HV9912 Preliminary Linear Dimming Linear dimming can be accomplished by varying the voltage at the IREF pin, as the output current is proportional to the voltage at the IREF pin. This can be done either by using a potentiometer from the REF pin or by applying an external voltage source at the IREF pin. In the HV9911, due to the offset voltage of the short circuit comparator as well as the non-linearity of the X2 gain stage, pulling the IREF pin very close to GND will cause the internal short circuit comparator to trigger and shut down the IC. To overcome this in the HV9912, the minimum output of the gain stage is limited to 125 ~ 250mV, allowing the IREF pin to be pulled all the way to 0V without triggering the short circuit comparator. Note: Since this control IC is a peak current mode controller, pulling the IREF pin to zero will not cause the LED current to become zero. The converter will still be operating at its minimum on-time causing a very small current to flow through the LEDs. To get zero LED current, the PWMD input has to be pulled to GND. LED string reconnects Output Cap Voltage 100V 90V 80V O VP O FF O VP ON LED string voltage LED string disconnects FLT t RC thiccup1 thiccup2 Output Current COMP 5V 1V Fig. 4 Open Circuit Protection 10 HV9912 Preliminary Pin Description Pin # 1 2 3 4 5 6 7 8 9 10 Pin VIN VDD GATE GND CS SC RT SYNC CLIM REF Description This pin is the input of a 90V high voltage regulator. This is a power supply pin for all internal circuits. It must be bypassed with a low ESR capacitor to GND (at least 0.1F). This pin is the output gate driver for an external N-channel power MOSFET. Ground return for all the low power analog internal circuitry. This pin must be connected to the return path from the input. This pin is used to sense the source current of the external power FET. It includes a built-in 100ns (min) blanking time. This pin is used to set the slope compensation. This pin sets the frequency of the power circuit. A resistor between RT and GND will program the circuit in constant frequency mode. This I/O pin may be connected to the SYNC pin of other HV9912 circuits and will cause the oscillators to lock to the highest frequency oscillator. This pin provides a programmable input current limit for the converter. The current limit can be set by using a resistor divider from the REF pin. This pin provides 2% accurate reference voltage. It must be bypassed with a 0.01F - 0.1F capacitor to GND. This pin is pulled to ground when there is an output short circuit condition or output over voltage condition. This pin can be used to drive an external MOSFET in the case of boost converters to disconnect the load from the source. This pin provides the over voltage protection for the converter. When the voltage at this pin exceeds 5V, the gate output of the HV9912 is turned off and FLT goes low. The IC will turn on when the voltage at the pin goes below 4.5V. When this pin is pulled to GND (or left open), switching of the HV9912 is disabled. When an external TTL high level is applied to it, switching will resume. Stable Closed loop control can be accomplished by connecting a compensation network between COMP and GND. This capacitor also controls the hiccup time. The voltage at this pin sets the output current level. The current reference can be set using a resistor divider from the REF pin. This pin provides output current feedback to the HV9912 by using a current sense resistor. 11 FAULT 12 13 14 15 16 OVP PWMD COMP IREF FDBK 11 HV9912 Preliminary 16-Lead SOIC (Narrow Body) Package Outline (NG) 9.90x3.90mm body, 1.75mm height (max), 1.27mm pitch 16 D 1 Note 1 (Index Area D/2 x E1/2) E1 E L2 Gauge Plane 1 L1 L Seating Plane Top View A View B View B h A A2 e Seating Plane h Note 1 A1 b A Side View View A-A Note 1: This chamfer feature is optional. If it is not present, then a Pin 1 identifier must be located in the index area indicated.The Pin 1 identifier may be either a mold, or an embedded metal or marked feature. Symbol MIN Dimension (mm) NOM MAX A 1.35 1.75 A1 0.10 0.25 A2 1.25 1.65 b 0.31 0.51 D 9.80 9.90 10.00 E 5.80 6.00 6.20 E1 3.80 3.90 4.00 e 1.27 BSC h 0.25 0.50 L 0.40 1.27 L1 1.04 REF L2 0.25 BSC 0 8 O 1 5O 15O O JEDEC Registration MS-012, Variation AC, Issue E, Sept. 2005. Drawings not to scale. (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to http://www.supertex.com/packaging.html.) Doc.# DSFP-HV9912 NR092607 12 |
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